Bandpass filter having parallel signal paths

ABSTRACT

A bandpass filter comprises a number of resonators which are arranged between an input and an output of the filter and which are interconnected to form at least two main signal paths that lead from the input to the output. The at least two main signal paths have overlapping passbands and are connected to the input and/or output via different resonators.

The present invention concerns a band pass filter for an electric orelectromagnetic signal, especially for a high-frequency signal. Suchfilters play an important role in the design of components for moderntelecommunications systems. The requirements that are generally imposedon such filters include steep filter flanks, high non-pass attenuation,uniform phase shift in the pass band region, etc. A distinction is madebetween different filter types, like Cauer, Tschebyscheff, Butterworthor Bessel filters, each of which satisfies one or more of theserequirements particularly well.

A common feature of all these filters is that they are constructed fromone or more resonators. In the simplest case of a filter with severalresonators, the individual resonators are connected in series, so that asingle signal path exists through the filter, on which a signal of allresonators pass through according to the sequence. The flank steepness,non-pass attenuation, etc., attainable with such an arrangement ofresonators, are established, among other things, by the number ofresonators.

Ordinary filter synthesis techniques additionally allow for thepossibility, in addition to a pure series circuit, of connectingindividual, not directly adjacent resonators of the filter to eachother, in order to produce overlapping of signal contributions in oneresonator, which can lead to zero setting of the transmission functionof the filter during finite arguments in the complex number plane.Filters synthesized with such methods always have a main signal paththat runs through all the resonators of the filter and, in addition tothe main signal path, one or more secondary signal paths that run fromthe input to the output of the filter via at least one coupling betweennon-adjacent resonators on the main signal path, and therefore have asmaller number of resonators in the main signal path.

Band pass filters are known from U.S. Pat. No. 6,337,610 B1 having twomain signal paths, i.e., two first signal paths, in which, unlike thesecondary signal paths of the ordinary filter structures, each has asecond signal path that passes through all the resonators of the firstpath in the same sequence, and one or more additional resonators betweenat least two directly consecutive resonators on the first path. Thesemain signal paths of these known filters each have a common input oroutput resonator connected to the input or output.

Practical implementation of such filters is connected with significantdemands and these demands are greater, the larger the number ofresonators n is, the more resonators are connected in series, and themore numerous are the secondary signal paths. Tuning conducted on aresonator can require corrections on adjacent resonators, those of themain signal path, and also possibly secondary signal paths, startingfrom the corresponding resonator. In the filters known from U.S. Pat.No. 6,337,610 B1, coupling between main signal paths is also possiblevia the common input and output resonators.

The task of the present invention is to provide a band pass filter,whose structure permits a simpler, faster and therefore morecost-effective filter implementation than the previous filterstructures.

The filter according to the invention is characterized by the fact thatthe several main signal paths that run from its input to its output haveno common resonators at the input and/or output, i.e., they areconnected via different resonators to the input and/or output. A changemade on one of the main signal paths can influence the behavior of theother main signal path, at best, via a single common resonator at theinput or output of the filter, and is therefore easy to handle in asimulation.

The main signal paths preferably have no common resonators either at theinput or output of the filter. Mutual influencing of the main signalpaths is then ruled out and they can be optimized fully independently ofeach other.

None of the main signal paths of the filter according to the inventionruns through all n resonators of the filter, so that a transmissionfunction can be assigned to each of these main signal paths, whichcorresponds to a smaller wave number than the total number n ofresonators. Amazingly, by overlapping of these transmission functions, atotal transmission function of the filter according to the inventionthat corresponds to an ordinary filter with a single main signal paththrough all end resonators is obtained. The advantage of the filterstructure according to the invention, however, is that its main signalpaths, because of the smaller wave number, can be implemented at lowercost that those of the ordinary filter, and that changes that are madeduring optimization on a resonator pertaining to only one of the mainsignal paths essentially affect only the transmission function of thismain signal path and leave the other main signal paths uninfluenced. Theproblem of implementing an n-pole filter can then be broken down intoimplementation of several partial filters corresponding to a main signalpath with a smaller wave number, these partial filters each having freeparameters that can be optimized without changing the transmissionfunctions of the other partial filters.

The filter structure according to the invention is applicable to anumber of filter types that are described below in conjunction with thefigures, with reference to practical examples.

FIGS. 1 a, 1 b show examples of structures of a filter according to theinvention with four resonators;

FIG. 2 shows, for comparison, the structure of an ordinary filter withfour resonators;

FIG. 3 shows the transmission and reflection function of a filter thatcan be implemented with the structure according to FIG. 1 a or accordingto FIG. 2;

FIG. 4 a, 4 b show coupling matrices for implementation of the filterwith the behavior depicted in FIG. 2 by means of the structure of FIG. 2a or FIG. 1 a;

FIG. 5 shows a schematic perspective of a filter according to theinvention with rectangular cavity resonators;

FIG. 6 shows a perspective, partially cutaway view of the filter withfour dielectrically loaded resonators;

FIGS. 7 a, 7 b show two sections through a first modification of thefilter from FIG. 6;

FIGS. 8 a, b show two sections through a second modification of a filterfrom FIG. 6;

FIG. 9 shows a perspective, partially cutaway view of a filter with fourcoaxial resonators;

FIG. 10 shows a view of a filter with four stripline resonators and thestructure according to FIG. 1 a;

FIG. 11 shows a schematic perspective view of a filter with a cavityresonator that uses higher wave types;

FIG. 12 shows a schematic view of the magnetic fields in the resonatorsof the filter from FIG. 10

FIGS. 1 a to 1 b each show a filter structure according to the inventionin comparison with the ordinary filter structure of FIG. 2.

In the ordinary filter structure, a signal path extends from input S ofthe filter to output L, passing through all four resonators 1 to 4 ofthe filter in series. The resonators 1 to 4 of the main signal path arestrongly coupled to each other, so that the comparatively weak directcoupling of the resonators 1 and 4 to each other via the secondarysignal path 5, depicted with a dashed line, during calculation of thebehavior of the filter can be treated as a disturbance in the filter,characterized essentially by the main signal path.

In contrast to this, in the filters of FIGS. 1 a, 1 b, there are no mainsignal paths, to which all the resonators belong. Instead, there are twomain signal paths that are formed, in the case of FIG. 1 a, byresonators 1, 2 or 3, 4 and, in the case of FIG. 1 b, by resonator 1 orresonators 2 to 4.

Since the main signal paths, in the case of FIGS. 1 a, 1 b, run from theinput S to the output L of the filter without any interaction with eachother, such a filter can be developed by initially calculating thecouplings into the individual main signal paths as a function of adesired transmission function of the entire filter, and thenimplementing the individual main signal paths completely independentlyof each other.

FIG. 3 shows the trend of the transmission characteristic, shown as asolid curve 8, and the reflection characteristic, shown as a dashedcurve 9, of a filter with four resonators. The characters 8, 9 areattainable with a filter having the structure depicted in FIG. 2 bymeans of the matrix of coupling coefficients depicted in FIG. 4 a. Theelements of the matrix that are situated on the positions directlyadjacent to the main diagonals correspond to the coupling coefficientsof the main signal path. Since all these positions have values differentfrom zero, the filter has precisely one main signal path. All elementsof the matrix that are not situated on either of these positions nor onthe main diagonals represent overcouplings of secondary signal paths. InFIG. 4 a, these are the elements 14 and 41, which describe a couplingwith resonators 1 and 4.

It is apparent that direct coupling between resonators 1 and 4 is muchsmaller than the coupling coefficients of the main signal path, so thatdirect coupling can be interpreted as a small correction of the signalmostly transmitted on the main signal path.

The trend of the transmission and reflection function as depicted inFIG. 3 is also attainable with the filter structure according to FIG. 1a, using the coupling matrix depicted in FIG. 4 b as a basis. It isapparent that the coupling coefficients of the two main signal paths S,1, 2, L and S, 3, 4, L have magnitudes of similar order, but in whichthe product of the coupling coefficients on signal path S, 1, 2, L ispositive, but, on the other hand, on signal path S, 3, 4, L it isnegative.

FIG. 5 shows a practical embodiment of a filter with the structuredepicted in FIG. 1 a. The input and output S and L are laid out asconnection parts 15 and 16 for a rectangular waveguide for transmissionof a microwave signal. In one end of the input connection part 15, twoiris diaphragms IS1, IS2 are formed, each of which discharges on acuboid resonator cavity 111 or 13, which embodies the resonator 11 or 13in FIG. 1 a. A microwave signal lying at the input connection part 15thus excites the H₁₀₁, wave type of the resonator cavities 11 and 13.The coupling coefficients between the input and resonators 1 and 3 areestablished by the configuration of the iris diaphragms IS1 and IS3. Inthe present case, the iris diaphragms IS1, IS3 extend from a broad side,on which the resonator cavities 11, 12 are opposite, from just abovehalf the height (in the y-direction) of the cavities and in the widthdirection (x-direction) centered roughly over half their width. Couplingof the two resonators 1, 3 to input S is therefore mostly inductive,which, by convention, can be equated to a coupling coefficient with apositive sign.

In an opposite end of the resonator cavities 11, 12, there are irisdiaphragms I12, I34, which discharge on cavities 12, 14, embodyingseries-connected resonators 2 and 4. The position and configuration ofthe iris diaphragm I12 corresponds to that of IS1, except for thedimensional differences reflecting the magnitude of the couplingcoefficient, so that coupling between resonators 1 and 2 is againinductive; on the other hand, the iris diaphragm I34 is slit-like andextends in the immediate vicinity of a side wall of the resonatorcavities 13, 14 over their entire width (in the x-direction) and iscapacitive on this account. A negative coupling coefficient betweenresonators 3, 4 is thus obtained.

Iris diaphragms I2L, I4L, which couple the resonator cavities 12, 14 tothe output connection 16, again have the same configuration as the irisdiaphragms IS1, IS3. Tunings of resonator frequencies of cavities 11 to14 that can be required because of different couplings between theresonators are achieved by tuning the widths of the cross sections orother tuning means known from prior art, for example, screws, pins, etc.

Since the two main signal paths S, 1, 2, L and S, 3, 4, L are fullyseparated from each other between the input and output connection, thecorresponding parts of the filter can be developed independently of eachother and tuned in production, in order to satisfy the correspondingrequirements of the coupling matrix. The connection of both main signalpaths at the input S and output L requires only slight corrections,since the interaction between the two is limited. The development andproduction are therefore reduced to implementation of two partialfilters, consisting of the resonators 1, 2 and 3, 4, which is muchsimpler than the usual development or tuning of a filter with fourseries-connected resonators, and the sensitivity of the behavior of afinished filter relative to manufacturing scatter also diminishes, sincethe effects of such scatter in a main signal path are essentiallyrestricted to it, and the second, or optionally other main signal pathsthat can be present in more complex filter structures than those shownhere are not affected detrimentally.

FIG. 6 shows the second practical example of the filter according to theinvention with the structure depicted schematically in FIG. 1 a. Ahousing 20 encloses an internal space that is divided by a partition 21arranged in the center with a cross-like layout into four chambers 22 to25 that form the four resonators 1, 2, 3, 4. In each chamber 22 to 25, adielectric element 26 is firmly attached to the bottom of the housingvia a spacer 27, and a tuning element 28 is mounted movable in the coverof housing 20 opposite dielectric element 26. The resonance frequency ofeach resonator is essentially determined by the dielectric element 26,in which any necessary fine tuning of the frequency is possible with thecorresponding tuning element 28. The spacer 27, like element 26,consists of a dielectric material, but with a much smaller dielectricconstant than element 26.

The input and output S, L of the filter are formed by coaxial linesections 30 and 31, whose external conductors 32 are each connected tohousing 20, whereas their internal conductor 33 is short-circuited tothe partition 21.

The coupling coefficients between the input S, the different resonators1, 2, 3, 4 and the output L are tunable by means of tuning screw 34, 35.Tuning screws 34, guided through the bottom of housing 20, near internalconductor 23, determine the coupling of input S to the resonators 1, 3.Screws arranged in the vicinity of output L in a mirror image of screws34 for tuning of the coupling between resonators 2 and 4 and output Lare covered and not visible in the figure. The tuning screws 35, whichare inserted into the side walls of housing 20 and, with their tips, lieopposite a transverse plate of the cross-like partition 21, serve fortuning the coupling between resonators 1 and 2 or between 3 and 4.

FIGS. 7 a, 7 b show a first modification of the filter from FIG. 6.Elements corresponding to each other are denoted with the same referencenumbers. The partition 21 between chambers 22 and 23 or 24 and 25 isenlarged, so that only a circular hole 29 remains as coupling openingbetween chambers 22, 23 or 24, 25. A metal wire 36 or 37 is passedthrough each of these holes 29 and connected on its two ends with theopposite surfaces of wall 21. The metal wires 36, 37 each produce a loopcoupling between the pairs of chambers operated as H₁₀₃ resonators.

The metal wire 36 is bent into a circle in a horizontal plane, and itstwo ends resting on wall 21 face each other. The metal wire 37, on theother hand, is bent S-shaped in the same horizontal plane; its two endsare supported on wall 21 on the sides of hole 29 facing away from eachother, through which it is guided. If we assume that the wave typesexcited in chambers 22, 24 are of the same phase, it is easy tocomprehend that, because of the different geometries of metal wires 36,37, magnetic fields with opposite direction or a phase differences of πcan be excited in chambers 23, 25, i.e., the coupling coefficientsbetween resonators 1, 2, on the one hand, and resonators 3, 4, on theother hand, have opposite signs.

A similar effect is achieved in the variants of FIGS. 9 a, 9 b. Thepartition 21 here is the same as in the variants of FIGS. 8 a, 8 b, buta metal wire 38 or 39 running through the holes 29 of partition 21 isnot connected on its ends to wall 21, but held in its hole 29 by adielectric element filling up hole 29 that passes throughelectromagnetic waves, and its ends freely extend into the chambers.

Whereas in wire 38, both free ends are deflected to the same side in thedirection of the longitudinal center plane of the filter, defined by theinternal conductor 33, those of the wire 39 are deflected to oppositesides. These two wires 38, 39 assure probe coupling between resonators1, 2 and 3, 4, each with opposite signs of the coupling coefficients.

FIG. 9 shows a third embodiment of a microwave filter with the structureof FIG. 1 a. Input S and output L of the filter are formed byrectangular waveguide sections 40, 41 with a height reduced incomparison with the connected waveguide sections 42. The waveguidesections 40, 41 forming the input and output are connected by two passbands 43, 44. Each of these pass bands 43, 44 includes two resonators 1,2 or 3, 4, each in the form of a resonator element 45, here cylindrical,galvanically connected and conducting with a bottom of the pass band 43,44, the elements being excitable to electrical oscillation by amicrowave signal lying at input S. The resonance frequency of eachresonator element 45 is established by its dimensions and the distanceto tuning screws 47 arranged in the upper wall 46 of the filter,opposite it. Tuning screws 47 are shown in FIG. 7 only for the resonatorelement 45 of pass band 43, but corresponding tuning screws (not shown)are also present for the resonator element 45 of pass band 44.

The pass band 43 between resonator element 45 is free, except for a tipof a tuning screw 48 that extends into the pass band, which serves fortuning the coupling between the two resonators of pass band 43. The passband 44 is blocked between these two resonator elements 45 on part ofits cross section by a partition 49. A tuning screw (not shown) that isarranged in the same manner as the tuning screw 48 depicted for the passband 43 in wall 46 and is opposite the upper edge of partition 49,permits tuning the coupling coefficient between resonators 3, 4 of passband 44.

Whereas the coupling between resonators 1, 2 of pass band 43 isinductive in nature, capacitive coupling of resonators 3, 4 is achievedby the partition 49 in pass band 44.

FIG. 10 shows the application of the principle according to theinvention to a filter in which resonators 1, 2, 3, 4 are formed by stripconductors 61 to 64 with length λ/2 structured on a substrate 60 inwhich λ is the wavelength of a signal propagating in the stripconductors in the pass band of the filter.

The strip conductor resonators 61, 62, 63, 64 are coupled to each otherand to an input conductor S and an output conductor L, extendingparallel and closely adjacent to each other over part of their length.In the main signal path formed by the strip conductors S, 61, 62, L, thestrip conductors 61, 62 are arranged so that the signal propagationdirection from input S to output L, each shown by arrows, is oriented inthe same direction in the sections of the strip conductors connected toeach other. In this way, the same sign of the coupling coefficient isobtained for all couplings on a main signal path S, 61, 62, L. Incontrast to this, on the main signal path S, 63, 64, L, the sections ofthe strip conductors 63, 64 connected to each other have oppositelyoriented signal propagation directions, so that a coupling coefficientwith a negative sign results between these two strip conductors.

Generally, the length of the strip conductor resonators can be nλ/2, inwhich n is a small natural number. When n is greater than 1, it is alsopossible to achieve different signs of the coupling coefficients on themain signal paths to produce couplings between the different half-wavesof the standing waves excited in the resonators, similar to thepractical example described below with reference to FIGS. 11 and 12.

FIGS. 11 and 12 show another practical example of the filter accordingto the invention, constructed like the practical example of FIG. 5 fromcavity resonators. This filter, shown in a perspective view of FIG. 12,includes only three resonators 2, 3, 4, which form two main signal paths2, 4 and 3, 4 with a common resonator 4. In the narrow side walls andends of the resonators 2, 3, 4, as well as the waveguide of the input Sand output L, diaphragms IS2, IS, I24, I34, I4L couple the resonators toeach other and to the input and output.

FIG. 12 shows the essential field distribution in the resonators in aschematic, sectional view. For the filter function, the H103 wave typeis utilized in the cavity resonators 2, 3, 4, shown in each case bymagnetic field lines in the resonators running in three closed circles.

The coupling coefficients on the individual iris diaphragms areestablished by their position relative to the field distribution in thecavities connecting them, as well as their cross section area. Thediaphragms IS2, IS3 each couple the left half-wave of input S in thesignal propagation direction (from left to right in FIG. 12) to thefirst half-wave of resonator 2 or 3. The magnetic fields of the firsthalf-waves excited in the resonators therefore have a direction ofrotation opposite to the last half-wave of input S, indicated by thearrows drawn on the circles.

The diaphragms I24 and I34 are laid out so that the first half-wave ofresonator 4 is coupled essentially to the third half-wave of resonator 3and the second half-wave of resonator 2, i.e., to half-waves withopposite sign. In this way, coupling coefficients with different signcan be obtained for coupling to diaphragm 134 and to diaphragm 124.

1-6. (canceled)
 7. A bandpass filter, comprising: a plurality ofresonators connected to each other between an input and an output of thefilter, and at least two main signal paths leading from the input to theoutput, the at least two main signal paths having overlapping passbandsand being connected via different ones of the resonators to the inputand the output.
 8. The bandpass filter according to claim 7, in that thetwo main signal paths have no common resonators.
 9. The bandpass filteraccording to claim 7, in that the at least two main signal paths haveoverall coupling coefficients with different signs.
 10. The bandpassfilter according to claim 7, in that at least one cavity resonator isunder the resonators.
 11. The bandpass filter according to claim 7, inthat at least one conduction resonator is under the resonators.
 12. Thebandpass filter according to claim 7, in that at least one striplineresonator is under the resonators.